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OPA846ID Datasheet(PDF) 11 Page - Texas Instruments |
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OPA846ID Datasheet(HTML) 11 Page - Texas Instruments |
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11 / 21 page  OPA846 11 SBOS250C www.ti.com WIDEBAND, HIGH-SENSITIVITY TRANSIMPEDANCE DESIGN The high GBP and low input voltage and current noise for the OPA846 make it an ideal wideband transimpedance ampli- fier. Very high transimpedance gains (> 100k Ω) benefit from the low input noise current of a JFET-input op amp, such as the OPA657. Unity-gain stability in the op amp is not required for application as a transimpedance amplifier. One transim- pedance design example is shown on the front page of this data sheet. Designs that require high bandwidths from a large area (high capacitance) detector with relatively low transimpedance gain will benefit from the low input voltage noise offered by the OPA846. This input voltage noise is peaked up over frequency at the output by the diode source capacitance, and can, in many cases, become the limiting factor to input sensitivity. The key elements of the design are the expected diode capacitance (CD) with the reverse bias voltage (–VB) applied, the desired transimpedance gain (RF), and the GBP of the OPA846 (1750MHz). Figure 3 shows a design using a 50pF detector diode capacitance and a 10k Ω transimpedance gain. With these three variables set (includ- ing the parasitic input capacitance for the OPA846 added to CD) the feedback capacitor (CF) value can be set to control the frequency response. To achieve a maximally flat 2nd- order Butterworth frequency response, set the feedback pole as shown in Equation 1. 1 24 ππ RC GBP RC FF FD = (1) The example of Figure 3 gives approximately 23MHz flat bandwidth using the 0.8pF feedback compensation. If the total output noise is bandlimited to a frequency less than the feedback pole frequency, a simple expression for the equiva- lent input noise current is given as Equation 3. II kT R E R EFC EQ N F N F ND =+ + + ( ) 2 2 2 4 2 3 π (3) Where: IEQ = equivalent input noise current if the output noise is bandlimited to F < 1/(2 πR FCF) IN = input current noise for the op amp inverting input EN = input voltage noise for the op amp CD = diode capacitance F = bandlimiting frequency in Hz (usually a post filter prior to further signal processing) 4kT = 1.6E – 20J at T = 290K Evaluating this expression up to the feedback pole frequency at 16.1MHz for the circuit of Figure 3 gives an equivalent input noise current of 4.9pA/ √Hz. This is much higher than the 2.8pA/ √Hz for just the op amp. This result is dominated by the last term in the equivalent input noise current calcu- lation from Equation 3. It is essential in this case to use a low- voltage noise op amp. For example, if a slightly higher input noise voltage, but otherwise identical op amp, was used instead of the OPA846 amplifier in this application noise amplifier (say 2.0nV/ √Hz), the total input-referred current noise would increase to 7.0pA/ √Hz. The output DC error for the circuit of Figure 3 is minimized by including the 10k Ω to ground on the noninverting input. This reduces the impact at the output of input bias current errors to the offset current times the feedback resistor. To minimize the output noise contribution of this resistor, a 0.01 µF capaci- tor is included in parallel. Worst-case output DC error for the circuit of Figure 3 at 25 °C is: VOS = ±0.6mV (input offset voltage) ± 0.35µA (input offset current) • 10k Ω = ±4.1mV Worst-case output offset DC drift is over the 0 °C to 70°C span is dVOS/dT = ±1.5µV/°C (input offset drift) ± 2nA/C (input offset current drift) • 10k Ω = ±21.5µV/°C Improved output DC precision and drift is possible, particu- larly at higher transimpedance gains, using the JFET input of the OPA657. The JFET input removes the input bias current from the error equation (eliminating the need for the resistor to ground on the noninverting input), leaving only the input offset voltage and drift as an output error term. Included in the characteristic curves are transimpedance frequency response curves for a fixed 10k Ω gain over vari- ous detector diode capacitance settings. These curves, along with the test circuit, are repeated in Figure 4. As the photo- FIGURE 3. Wideband, Low Noise, Transimpedance Amplifier. Adding the common-mode and differential-mode input capaci- tance (1.8 + 2.0)pF to the 50pF diode source capacitance of Figure 3, with a 10k Ω transimpedance gain using the 1750MHz GBP for the OPA846, requires a feedback pole set to 16.1MHz. This requires a 1pF total feedback capacitance. Typical sur- face-mount resistors have 0.2pF parasitic capacitance leaving a required extrinsic 0.8pF value, as shown in Figure 3. Equation 2 gives the approximate –3dB bandwidth, if CF is set using Equation 1. f GBP RC Hz dB FD − = ( ) 3 2 π (2) R F 10k Ω +5V –5V C D 50pF λ 0.01 µF 10k Ω OPA846 –V B I D V O = ID RF Power-supply decoupling not shown. C F 0.8pF |
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