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ADN2850BRUZ25 Datasheet(PDF) 23 Page - Analog Devices |
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ADN2850BRUZ25 Datasheet(HTML) 23 Page - Analog Devices |
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23 / 30 page  Data Sheet ADN2850 Rev. F | Page 23 of 30 APPLICATIONS INFORMATION GAIN CONTROL COMPENSATION A digital resistor is commonly used in gain control such as the noninverting gain amplifier shown in Figure 34. U1 VO R2 250k Ω VI R1 47k Ω C1 11pF W B C2 2.2pF Figure 34. Typical Noninverting Gain Amplifier When the RDAC B terminal parasitic capacitance is connected to the op amp noninverting node, it introduces a zero for the 1/βO term with 20 dB/dec, whereas a typical op amp gain bandwidth product (GBP) has −20 dB/dec characteristics. A large R2 and finite C1 can cause the frequency of this zero to fall well below the crossover frequency. Therefore, the rate of closure becomes 40 dB/dec, and the system has a 0° phase margin at the crossover frequency. If an input is a rectangular pulse or step function, the output can ring or oscillate. Similarly, it is also likely to ring when switching between two gain values; this is equivalent to a stop change at the input. Depending on the op amp GBP, reducing the feedback resistor might extend the frequency of the zero far enough to overcome the problem. A better approach is to include a compensation capacitor, C2, to cancel the effect caused by C1. Optimum compensation occurs when R1 × C1 = R2 × C2. This is not an option because of the variation of R2. As a result, one can use the previous relationship and scale C2 as if R2 were at its maximum value. Doing this might overcompensate and compromise the performance when R2 is set at low values. Alternatively, it avoids the ringing or oscillation at the worst case. For critical applications, find C2 empirically to suit the oscillation. In general, C2 in the range of a few picofarads to no more than a few tenths of picofarads is usually adequate for the compensation. Similarly, W and A terminal capacitances are connected to the output (not shown); their effect at this node is less significant and the compensation can be avoided in most cases. PROGRAMMABLE LOW-PASS FILTER In analog-to-digital conversions (ADCs), it is common to include an antialiasing filter to band limit the sampling signal. Therefore, the dual-channel ADN2850 can be used to construct a second-order Sallen-Key low-pass filter, as shown in Figure 35. B VI AD8601 +2.5V VO ADJUSTED CONCURRENTLY –2.5V V+ V– W R R2 R1 B W R C1 C2 U1 Figure 35. Sallen-Key Low-Pass Filter The design equations are 2 2 2 f f f I O S Q S V V ω ω ω (10) C2 C1 R2 R1 O 1 ω (11) Q = C2 R2 1 C1 R1 1 (12) First, users should select convenient values for the capacitors. To achieve maximally flat bandwidth, where Q = 0.707, let C1 be twice the size of C2 and let R1 equal R2. As a result, the user can adjust R1 and R2 concurrently to the same setting to achieve the desirable bandwidth. PROGRAMMABLE OSCILLATOR In a classic Wien bridge oscillator, the Wien network (R||C, R'C') provides positive feedback, whereas R1 and R2 provide negative feedback (see Figure 36). D1 D2 OP1177 V+ V– +2.5V + – –2.5V VO U1 R2A 2.1k Ω R2B 10k Ω B A W R1 1k Ω AMPLITUDE ADJUSTMENT R = R' = ADN2850 R2B = AD5231 D1 = D2 = 1N4148 R' 25k Ω A B W C' VP R 25k Ω B W C 2.2nF FREQUENCY ADJUSTMENT 2.2nF Figure 36. Programmable Oscillator with Amplitude Control |
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