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NCV887001D1R2G Datasheet(PDF) 9 Page  ON Semiconductor 

NCV887001D1R2G Datasheet(HTML) 9 Page  ON Semiconductor 
9 / 16 page NCV8870 www.onsemi.com 9 Both duty cycles will actually be higher due to power loss in the conversion. The exact duty cycles will depend on conduction and switching losses. If the maximum input voltage is higher than the output voltage, the minimum duty cycle will be negative. This is because a boost converter cannot have an output lower than the input. In situations where the input is higher than the output, the output will follow the input, minus the diode drop of the output diode and the converter will not attempt to switch. If the calculated Dmax is higher the Dmax of the NCV8870, the conversion will not be possible. It is important for a boost converter to have a restricted Dmax, because while the ideal conversion ration of a boost converter goes up to infinity as D approaches 1, a real converter’s conversion ratio starts to decrease as losses overtake the increased power transfer. If the converter is in this range it will not be able to regulate properly. If the following equation is not satisfied, the device will skip pulses at high VIN: D min fs w t on(min) Where: fs: switching frequency [Hz] ton(min): minimum on time [s] 2. Select Current Sense Resistor Current sensing for peak current mode control and current limit relies on the MOSFET current signal, which is measured with a ground referenced amplifier. The easiest method of generating this signal is to use a current sense resistor from the source of the MOSFET to device ground. The sense resistor should be selected as follows: R S + V CL I CL Where: RS: sense resistor [ W] VCL: current limit threshold voltage [V] ICL: desire current limit [A] 3. Select Output Inductor The output inductor controls the current ripple that occurs over a switching period. A high current ripple will result in excessive power loss and ripple current requirements. A low current ripple will result in a poor control signal and a slow current slew rate in case of load steps. A good starting point for peak to peak ripple is around 20−40% of the inductor current at the maximum load at the worst case VIN, but operation should be verified empirically. The worst case VIN is half of VOUT, or whatever VIN is closest to half of VOUT. After choosing a peak current ripple value, calculate the inductor value as follows: L + V IN(WC) DWC DI L,max fs Where: VIN(WC): VIN value as close as possible to half of VOUT [V] DWC: duty cycle at VIN(WC) DIL,max: maximum peak to peak ripple [A] The maximum average inductor current can be calculated as follows: I L,AVG + V OUTIOUT(max) V IN(min)h The Peak Inductor current can be calculated as follows: I L,peak + IL,avg ) DI L,max 2 Where: IL,peak: Peak inductor current value [A] 4. Select Output Capacitors The output capacitors smooth the output voltage and reduce the overshoot and undershoot associated with line transients. The steady state output ripple associated with the output capacitors can be calculated as follows: DI OUT(max) fC OUT ) I OUT(max) 1 * D ) V IN(min)D 2fL R ESR V OUT(ripple) + The capacitors need to survive an RMS ripple current as follows: I Cout(RMS) + IOUT D WC D WC ) D WC 12 D WC L R OUT T SW 2 The use of parallel ceramic bypass capacitors is strongly encouraged to help with the transient response. 5. Select Input Capacitors The input capacitor reduces voltage ripple on the input to the module associated with the ac component of the input current. I Cin(RMS) + V IN(min) 2 D WC LfsVOUT23 6. Select Feedback Resistors The feedback resistors form a resistor divider from the output of the converter to ground, with a tap to the feedback pin. During regulation, the divided voltage will equal Vref. The lower feedback resistor can be chosen, and the upper feedback resistor value is calculated as follows: Rupper + Rlower Vout * Vref V ref The total feedback resistance (Rupper + Rlower) should be in the range of 1 k W – 100 kW. 
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