PRODUCT SPECIFICATION

ML4827

REV. 1.0.1 6/27/01

9

Error Amplifier Compensation

The PWM loading of the PFC can be modeled as a negative

resistor; an increase in input voltage to the PWM causes a

decrease in the input current. This response dictates the

proper compensation of the two transconductance error

amplifiers. Figure 2 shows the types of compensation net-

works most commonly used for the voltage and current error

amplifiers, along with their respective return points. The cur-

start characteristic on the PFC: as the reference voltage

comes up from zero volts, it creates a differentiated voltage

on IEAO which prevents the PFC from immediately

demanding a full duty cycle on its boost converter.

There are two major concerns when compensating the volt-

age loop error amplifier; stability and transient response.

Optimizing interaction between transient response and sta-

bility requires that the error amplifier’s open-loop crossover

frequency should be 1/2 that of the line frequency, or 23Hz

for a 47Hz line (lowest anticipated international power fre-

quency). The gain vs. input voltage of the ML4827’s voltage

error amplifier has a specially shaped nonlinearity such that

under steady-state operating conditions the transconductance

of the error amplifier is at a local minimum. Rapid perturba-

tions in line or load conditions will cause the input to the

nal) value. If this happens, the transconductance of the volt-

age error amplifier will increase significantly, as shown in the

Typical Performance Characteristics. This raises the gain-

bandwidth product of the voltage loop, resulting in a much

more rapid voltage loop response to such perturbations than

would occur with a conventional linear gain characteristic.

The current amplifier compensation is similar to that of the

voltage error amplifier with the exception of the choice of

crossover frequency. The crossover frequency of the current

amplifier should be at least 10 times that of the voltage

amplifier, to prevent interaction with the voltage loop. It

should also be limited to less than 1/6th that of the switching

frequency, e.g. 16.7kHz for a 100kHz switching frequency.

There is a modest degree of gain contouring applied to the

transfer characteristic of the current error amplifier, to

increase its speed of response to current-loop perturbations.

However, the boost inductor will usually be the dominant

factor in overall current loop response. Therefore, this con-

touring is significantly less marked than that of the voltage

error amplifier.

For more information on compensating the current and volt-

age control loops, see Application Notes 33 and 34. Applica-

tion Note 16 also contains valuable information for the

design of this class of PFC.

Oscillator (RAMP 1)

lator output clock:

The deadtime of the oscillator is derived from the following

equation:

The deadtime of the oscillator may be determined using:

ating frequency can typically be approximated by:

EXAMPLE:

For the application circuit shown in the data sheet, with the

oscillator running at:

Ω.

The deadtime of the oscillator adds to the Maximum PWM

Duty Cycle (it is an input to the Duty Cycle Limiter). With

zero oscillator deadtime, the Maximum PWM Duty Cycle is

typically 45% for the ML4827-1. In many applications of the

large as to extend the Maximum Duty Cycle beyond 50%.

This can be accomplished by using a stable 470pF capacitor

f

OSC

1

t

RAMP

t

DEADTIME

+

---------------------------------------------------

=

(2)

t

RAMP

C

T

=

R

T

×

In

V

REF

1.25

–

V

REF

3.75

–

--------------------------------





×

(3)

t

RAMP

C

T

=

R

T

×

0.51

×

t

DEADTIME

2.5V

5.1mA

------------------

=

C

T

×

490

=

C

T

×

(4)

f

OSC

1

t

RAMP

----------------

=

(5)

f

OSC

100kHz

1

t

RAMP

----------------

==

t

RAMP

C

T

=

R

T

×

0.51

×

1

=

10

5

–

×